Magnetics designer

Pick an inductor or a flyback transformer, a core and the windings. The two table cores carry the values TDK publishes for them in its N87 ferrite [TDK Electronics, E 25/13/7 (EF 25)], [TDK Electronics, ETD 29/16/10]. For any other core, choose “Other core” and enter the values from its data sheet. The designer gives:

• the fewest turns that keep the peak flux density within a limit, and the air gap that then gives the inductance;
• for a table core, the gapped core sets its data sheet offers, with the turns each one needs;
• the peak flux density, and the ac flux amplitude that the core loss depends on;
• the window utilization, and each winding’s dc resistance and ac resistance factor at the switching frequency;
• for a flyback, the leakage inductance of the winding arrangement.

Each result names its equation. The chart shows each winding’s ac resistance factor against frequency.

What it computes

Enter the inductance and the peak and rms current of the winding. For a flyback these are the magnetizing inductance referred to the primary and the primary’s currents, and the secondary’s rms current is entered separately. The converter designer gives the inductance and the peak current. The ripple $\Delta i$ is half the peak-to-peak ripple, as everywhere on this site. $B_\mathrm{max}$ is the largest flux density the design allows: below the material’s saturation flux density at the core’s highest temperature, with a margin [Erickson & Maksimović 2020, Ch. 11].

Turns

The flux density follows the current (peak flux density). Holding its peak at $B_\mathrm{max}$ gives the fewest turns [Erickson & Maksimović 2020, Ch. 11]:

$$N = \frac{L I_\mathrm{pk}}{B_\mathrm{max} A_e}$$

$N$

number of turns

$A_e$

effective core area

$B_\mathrm{max}$

largest flux density allowed

$I_\mathrm{pk}$

peak current

$L$

inductance

Fewest turns that keep the peak flux density at a limit✓ verified by 7 tests

Assumptions, convention and source

Assumptions

• linear (unsaturated) magnetic material
• flux density uniform over A_e

Notes

more turns lower the peak flux density at the same inductance and current, but lengthen the winding and raise its resistance

Source

Erickson & Maksimović 2020, Ch. 11 (Inductor Design); derived here

Derivation

step-by-step sympy derivation

Equation id

mag.N_Bmax · equations.yaml entry

The tool evaluates it at the core’s smallest cross-section $A_\mathrm{min}$, where the flux density is highest, and rounds up to whole turns. More turns lower the flux density further, but lengthen the winding and raise its resistance. Enter $N$ to use more.

The gap

With the turns fixed, the inductance sets the gap. The magnetic circuit of a gapped core (inductance factor of a gapped core), solved for the gap length, gives:

$$l_g \approx \frac{\mu_0 A_e N^{2}}{L} - \frac{l_e}{\mu_i}$$

$l_g$

air-gap length

$A_e$

effective core area

$L$

inductance

$N$

number of turns

$l_e$

effective magnetic path length

$\mu_i$

initial relative permeability

$\mu_0$

permeability of free space

Air-gap length that gives an inductance with N turns✓ verified by 7 tests

Assumptions, convention and source

Assumptions

• linear (unsaturated) magnetic material
• no fringing flux at the gap
• flux density uniform over A_e

Notes

a negative result means the ungapped core already gives less than this inductance with N turns: more turns are needed; fringing lets a real gap be somewhat longer than this estimate

Source

Erickson & Maksimović 2020, Ch. 11 (Inductor Design); derived here

Derivation

step-by-step sympy derivation

Equation id

mag.gap_length · equations.yaml entry

Here $\mu_i$ is the effective permeability of the ungapped set. The tool takes it from the data sheet’s $A_L$ of the ungapped set, through the same equation with no gap. A negative result means that even the ungapped core gives less than the inductance with these turns: add turns.

This estimate ignores fringing. Around the gap the flux spreads beyond the core’s area, so a real gap passes more flux than the estimate assumes, and it must be longer for the same inductance. A data sheet therefore gives the $A_L$ of its gapped core sets, each a gapped core with an ungapped one [TDK Electronics, E 25/13/7 (EF 25)]. For a table core the tool lists these sets. For each it shows the turns that give the inductance, the inductance of those whole turns and the peak flux density at $A_\mathrm{min}$. The button in a row uses its turns. Use a set with a large gap, and compare its gap with the tool’s estimate for those turns: the real gap is clearly longer.

Flux density and its swing

The peak flux density at $A_\mathrm{min}$ is what must stay below saturation. The core loss depends on the swing instead. A ripple of $\pm\Delta i$ moves the flux density about its dc value by:

$$B_\mathrm{ac} = \frac{L \Delta i_L}{N A_e}$$

$B_\mathrm{ac}$

ac flux-density amplitude

$A_e$

effective core area

$\Delta i_L$

inductor ripple, half peak-to-peak

$L$

inductance

$N$

number of turns

AC flux-density amplitude from the current ripple✓ verified by 7 tests

Assumptions, convention and source

Assumptions

• linear (unsaturated) magnetic material
• flux density uniform over A_e

Convention

$\Delta i_L$ is half the peak-to-peak ripple (Erickson), so $B_\mathrm{ac}$ is half the peak-to-peak swing $\Delta B$: the amplitude that core-loss data take

Notes

the flux density follows the current, so the ripple alone sets its swing, the quantity core-loss data are given for; the dc current sets where the swing lies, and with it the margin to saturation ($B_\mathrm{pk}$)

Source

Erickson & Maksimović 2020, Ch. 10 (Basic Magnetics Theory); derived here

Derivation

step-by-step sympy derivation

Equation id

mag.B_ac · equations.yaml entry

This is the amplitude that core-loss data take (core loss per volume): read the loss per volume off the material’s loss curves at $B_\mathrm{ac}$ and $f_s$, or use the loss budget. In a CCM inductor the ripple is a small part of the current, and so is $B_\mathrm{ac}$ of $B_\mathrm{pk}$. In a DCM flyback the current falls to zero every period, and the swing is the whole peak.

Copper and the window

Each winding is round wire: $k_s$ strands in parallel, each of bare diameter $d_w$ and diameter $d_o$ over its insulation, wound in $M_\ell$ layers across the coil former’s winding breadth $b_w$. The copper of one turn, and the part of the window that all windings fill:

$$A_w = \frac{k_s \pi d_w^{2}}{4}$$

$A_w$

bare conductor area of one turn

$d_w$

bare wire diameter

$k_s$

strands in parallel

Copper area of one turn of round strands✓ verified by 8 tests

Assumptions, convention and source

Assumptions

• definition

Notes

the bare diameter, without the insulation; wire tables give both

Source

National Bureau of Standards 1966

Equation id

wind.round_area · equations.yaml entry

$$K_u = \frac{N A_w}{W_A}$$

$K_u$

window utilization

$A_w$

bare conductor area of one turn

$N$

number of turns

$W_A$

winding window area

Window utilization of a winding✓ verified by 7 tests

Assumptions, convention and source

Assumptions

• definition

Notes

insulation, the bobbin and the gaps between round wires leave only part of the window to copper; a design keeps K_u below what it can wind

Source

Erickson & Maksimović 2020, Ch. 11 (Inductor Design)

Equation id

wind.fill · equations.yaml entry

$K_u$ must stay below what can actually be wound. Insulation, the coil former and the gaps between round wires take the rest [Erickson & Maksimović 2020, Ch. 11]. The tool warns above $K_{u,\mathrm{max}}$.

DC resistance

Copper’s resistivity rises with temperature. For annealed copper the tool uses the values of the NBS copper wire tables [National Bureau of Standards 1966]:

$$\rho_w = \rho_{20} \left(1 + \alpha_{20} \left(T_w - 20\right)\right)$$

$\rho_w$

conductor resistivity

$T_w$

winding temperature

$\alpha_{20}$

temperature coefficient at 20 °C

$\rho_{20}$

resistivity at 20 °C

Resistivity of a conductor at its temperature✓ verified by 8 tests

Assumptions, convention and source

Assumptions

• resistance linear in temperature near 20 °C

Notes

for annealed copper $\rho_{20} = 1.7241 \times 10^{-8}\,\Omega\,\mathrm{m}$ and $\alpha_{20} = 0.00393$ per kelvin; a winding at 100 °C has about 31 % more resistance than at 20 °C

Source

National Bureau of Standards 1966

Equation id

wind.rho_T · equations.yaml entry

$$R_\mathrm{dc} = \frac{\rho_w N \mathrm{MLT}}{A_w}$$

$R_\mathrm{dc}$

winding dc resistance

$A_w$

bare conductor area of one turn

$\mathrm{MLT}$

mean length per turn

$N$

number of turns

$\rho_w$

conductor resistivity

DC resistance of a winding✓ verified by 7 tests

Assumptions, convention and source

Assumptions

• a dc current spread uniformly over the conductor's cross-section

Notes

the winding is a conductor $N \cdot \mathrm{MLT}$ long; the resistivity of copper rises with temperature

Source

Erickson & Maksimović 2020, Ch. 11 (Inductor Design); derived here

Derivation

step-by-step sympy derivation

Equation id

wind.dcr · equations.yaml entry

$I_\mathrm{rms}^2 R_\mathrm{dc}$ (conduction loss) is the loss the current would cause if it were dc. The ac part of the current adds to it, as the next section shows.

AC resistance (Dowell)

At high frequency the current crowds into a surface layer about a skin depth thick [Erickson & Maksimović 2020, Ch. 10]:

$$\delta = \sqrt{\frac{\rho_w}{\pi \mu_0 f}}$$

$\delta$

skin depth

$f$

frequency

$\rho_w$

conductor resistivity

$\mu_0$

permeability of free space

Skin depth of a conductor✓ verified by 8 tests

Assumptions, convention and source

Assumptions

• a good conductor carrying a sinusoidal current (displacement current neglected)

Notes

the current density falls by a factor $e$ for every skin depth into the conductor; conductors much thicker than $\delta$ carry the current in a surface layer

Source

Erickson & Maksimović 2020, Ch. 10 (Basic Magnetics Theory: eddy currents in winding conductors); derived here

Derivation

step-by-step sympy derivation

Equation id

wind.skin_depth · equations.yaml entry

In a winding of several layers, each layer also carries the field of the layers below it, which induces eddy currents in it: the proximity effect. Dowell’s analysis treats each layer as a foil [Dowell 1966]. For round wire, each strand becomes a square of the same area, and the layer becomes a foil of that thickness whose gaps lower its conductivity [Erickson & Maksimović 2020, Ch. 10]:

$$\eta = \frac{N_\ell k_s \sqrt{\frac{\pi}{4}} d_w}{b_w}$$

$\eta$

porosity of a layer

$N_\ell$

turns per layer

$b_w$

winding breadth

$d_w$

bare wire diameter

$k_s$

strands in parallel

Porosity of a layer of round conductors✓ verified by 8 tests

Assumptions, convention and source

Assumptions

• each round conductor replaced by a square conductor of the same area, the layer by a foil of that thickness

Notes

a square of side $\sqrt{\pi/4}\,d_w$ has the strand's area; a layer whose squares fill the whole breadth has porosity 1, and a value above 1 means the turns do not fit in the layer

Source

Dowell 1966; Erickson & Maksimović 2020, Ch. 10 (Basic Magnetics Theory: foil windings and layers)

Equation id

wind.porosity · equations.yaml entry

$$\varphi = \frac{\sqrt{\eta} \sqrt{\frac{\pi}{4}} d_w}{\delta}$$

$\varphi$

layer thickness over skin depth

$d_w$

bare wire diameter

$\delta$

skin depth

$\eta$

porosity of a layer

Dowell's layer thickness for round conductors✓ verified by 8 tests

Assumptions, convention and source

Assumptions

• each round conductor replaced by a square conductor of the same area, the layer by a foil of that thickness

Notes

the equivalent foil is $\sqrt{\pi/4}\,d_w$ thick; its gaps lower its effective conductivity by the porosity, which enters as $\sqrt{\eta}$

Source

Dowell 1966; Erickson & Maksimović 2020, Ch. 10 (Basic Magnetics Theory: foil windings and layers)

Equation id

wind.phi_round · equations.yaml entry

$$F_R = \varphi \left(\frac{\sinh{\left(2 \varphi \right)} + \sin{\left(2 \varphi \right)}}{\cosh{\left(2 \varphi \right)} - \cos{\left(2 \varphi \right)}} + \frac{2 \left(M_\ell^{2} - 1\right) \left(\sinh{\left(\varphi \right)} - \sin{\left(\varphi \right)}\right)}{3 \left(\cosh{\left(\varphi \right)} + \cos{\left(\varphi \right)}\right)}\right)$$

$F_R$

ac resistance factor

$M_\ell$

layers from zero to peak MMF

$\varphi$

layer thickness over skin depth

AC resistance factor of a foil winding (Dowell)✓ verified by 9 tests

Assumptions, convention and source

Assumptions

• foil layers spanning the winding breadth, a sinusoidal current, and a one-dimensional field in the window

Notes

the first term is the skin effect in each layer, the second the proximity effect of the layers' fields on each other, growing as $M_\ell^2$; with $\varphi$ well below 1 the factor tends to 1

Source

Dowell 1966; Erickson & Maksimović 2020, Ch. 10 (Basic Magnetics Theory: eddy currents in winding conductors)

Equation id

wind.dowell · equations.yaml entry

$F_R$ is the ratio of the ac resistance to $R_\mathrm{dc}$ at one frequency. Each harmonic of the current sees the factor at its own frequency, so the chart plots $F_R$ against frequency, with $f_s$ marked. With $\varphi$ above 1, each added layer raises $F_R$ steeply, as $M_\ell^2$. The remedies are fewer layers of thinner strands in parallel, or a foil.

In an inductor with a small ripple, the ac part of the current is small and $F_R$ matters little: the table’s dc loss is then most of the copper loss. A flyback’s windings carry pulses, so much of each winding’s rms current is ac. Its harmonics see $F_R$ at their frequencies, and the copper loss can be several times the dc loss.

Leakage inductance (flyback)

The field between the windings stores energy that does not couple them: the leakage inductance. With the secondary wound over the primary (P-S):

$$L_\mathrm{lk} = \frac{\mu_0 N^{2} \mathrm{MLT} \left(\frac{h_p}{3} + h_g + \frac{h_s}{3}\right)}{b_w}$$

$L_\mathrm{lk}$

leakage inductance (primary)

$\mathrm{MLT}$

mean length per turn

$N$

number of turns

$b_w$

winding breadth

$h_g$

spacing between the windings

$h_p$

primary winding height

$h_s$

secondary winding height

$\mu_0$

permeability of free space

Leakage inductance of a primary-secondary winding arrangement✓ verified by 7 tests

Assumptions, convention and source

Assumptions

• one-dimensional field in the winding window: the MMF changes only across the layers, and the field runs along the winding breadth

Notes

referred to the primary; each winding counts with a third of its height and the spacing in full, and a broad, low window gives the least leakage

Source

Erickson & Maksimović 2020, Ch. 10 (Basic Magnetics Theory: leakage flux in windings, MMF diagrams); derived here

Derivation

step-by-step sympy derivation

Equation id

xfmr.leakage.ps · equations.yaml entry

Splitting the primary into two halves around the secondary (P-S-P) halves the peak magnetomotive force in the window, and the leakage falls to about a quarter [Erickson & Maksimović 2020, Ch. 10]:

$$L_\mathrm{lk} = \frac{\mu_0 N^{2} \mathrm{MLT} \left(\frac{h_p}{3} + 2 h_g + \frac{h_s}{3}\right)}{4 b_w}$$

$L_\mathrm{lk}$

leakage inductance (primary)

$\mathrm{MLT}$

mean length per turn

$N$

number of turns

$b_w$

winding breadth

$h_g$

spacing between the windings

$h_p$

primary winding height

$h_s$

secondary winding height

$\mu_0$

permeability of free space

Leakage inductance with the primary split around the secondary (P-S-P)✓ verified by 7 tests

Assumptions, convention and source

Assumptions

• one-dimensional field in the winding window: the MMF changes only across the layers, and the field runs along the winding breadth

Notes

splitting the primary halves the peak MMF, so the stored energy, and with it the leakage, falls to about a quarter

Source

Erickson & Maksimović 2020, Ch. 10 (Basic Magnetics Theory: interleaving the windings); derived here

Derivation

step-by-step sympy derivation

Equation id

xfmr.leakage.psp · equations.yaml entry

The tool takes a winding’s height as its layers times $d_o$, and $h_g$ as the insulation between two windings (for P-S-P, at each of the two interfaces). In a flyback the leakage energy goes to the clamp: take this inductance to the clamp check.

Worked examples

A buck inductor on the E 25/13/7 set:

Buck inductor example

Symbol	Value	How it is obtained
$L$ inductance	100 µH	given
$I_{L,\mathrm{pk}}$ peak inductor current	2.2 A	given
$I_{L,\mathrm{rms}}$ rms inductor current	2 A	given
$\Delta i_L$ inductor ripple, half peak-to-peak	200 mA	given
$f_s$ switching frequency	100 kHz	given
$B_\mathrm{max}$ largest flux density allowed	300 mT	given
$T_w$ winding temperature	100 °C	given
$b_w$ winding breadth	15 mm	given
$K_{u,\mathrm{max}}$ largest window utilization that can be wound	0.4	given
$d_w$ bare wire diameter	0.8 mm	given
$d_o$ wire diameter over its insulation	0.86 mm	given
$k_s$ strands in parallel	1	given
$M_\ell$ layers of the winding (Dowell's M)	1	given
$A_e$ effective core area	52.5 mm²	TDK Electronics, E 25/13/7 (EF 25)
$A_\mathrm{min}$ smallest cross-section of the core	51.5 mm²	TDK Electronics, E 25/13/7 (EF 25)
$l_e$ effective magnetic path length	57.5 mm	TDK Electronics, E 25/13/7 (EF 25)
$A_{L0}$ inductance factor of the ungapped set	1.85 µH	TDK Electronics, E 25/13/7 (EF 25)
$W_A$ winding window area	61 mm²	TDK Electronics, E 25/13/7 (EF 25)
$\mathrm{MLT}$ mean length per turn	50 mm	TDK Electronics, E 25/13/7 (EF 25)
$N_\mathrm{min}$ fewest turns for Bmax, at Amin	14.24	Fewest turns that keep the peak flux density at a limit
$N$ number of turns	15	rounded up
$A_L$ inductance factor the design needs	444.4 nH	Inductance from the inductance factor
$\mu_e$ effective permeability of the ungapped set	1612	Inductance factor of a gapped core
$l_g$ gap, without fringing	0.1128 mm	Air-gap length that gives an inductance with N turns
$B_\mathrm{pk}$ peak flux density (at Amin)	284.8 mT	Peak flux density from inductance and peak current
$B_\mathrm{ac}$ ac flux-density amplitude	25.4 mT	AC flux-density amplitude from the current ripple
$\rho_w$ conductor resistivity (at Tw)	22.66 nΩ·m	Resistivity of a conductor at its temperature
$\delta$ skin depth (at fs)	0.2396 mm	Skin depth of a conductor
$A_w$ bare conductor area of one turn	0.5027 mm²	Copper area of one turn of round strands
$R_\mathrm{dc}$ winding dc resistance	33.81 mΩ	DC resistance of a winding
$\eta$ porosity of a layer	0.709	Porosity of a layer of round conductors
$\varphi$ layer thickness over skin depth	2.492	Dowell's layer thickness for round conductors
$F_R$ ac resistance factor (at fs)	2.468	AC resistance factor of a foil winding (Dowell)
$P_\mathrm{dc}$ loss at the dc resistance	135.3 mW	Conduction loss in a resistance
$K_u$ window utilization	0.1236	Window utilization of a winding

The fewest turns need a gap that no set of the data sheet has. In the tool’s table of gapped sets, the smallest gap needs as many turns but gives more inductance, so its peak flux density exceeds $B_\mathrm{max}$. The next set stays within it with a few more turns. One layer of this wire has an $F_R$ well above 1 at the switching frequency, from the skin effect alone, but only the small ripple sees it.

A flyback transformer on the ETD 29/16/10 set, wound P-S. Its operating point is the converter designer’s flyback example at its lowest input voltage, with the inductance set to its ripple target:

Flyback transformer example

Symbol	Value	How it is obtained
$L_M$ magnetizing inductance (primary)	600 µH	given
$I_{p,\mathrm{pk}}$ peak primary current	1.37 A	given
$I_{p,\mathrm{rms}}$ rms primary current	914 mA	given
$\Delta i_M$ magnetizing ripple, half peak-to-peak	174 mA	given
$f_s$ switching frequency	100 kHz	given
$B_\mathrm{max}$ largest flux density allowed	300 mT	given
$n$ turns ratio, secondary/primary	0.25	given
$I_{s,\mathrm{rms}}$ rms secondary current	3.1 A	given
$T_w$ winding temperature	100 °C	given
$b_w$ winding breadth	19 mm	given
$K_{u,\mathrm{max}}$ largest window utilization that can be wound	0.4	given
$h_g$ spacing between the windings	0.1 mm	given
$d_w$ bare wire diameter (primary)	0.5 mm	given
$d_o$ wire diameter over its insulation (primary)	0.56 mm	given
$k_s$ strands in parallel (primary)	1	given
$M_\ell$ layers of the winding (Dowell's M) (primary)	2	given
$d_w$ bare wire diameter (secondary)	0.63 mm	given
$d_o$ wire diameter over its insulation (secondary)	0.69 mm	given
$k_s$ strands in parallel (secondary)	2	given
$M_\ell$ layers of the winding (Dowell's M) (secondary)	1	given
$A_e$ effective core area	76 mm²	TDK Electronics, ETD 29/16/10
$A_\mathrm{min}$ smallest cross-section of the core	71 mm²	TDK Electronics, ETD 29/16/10
$l_e$ effective magnetic path length	70.4 mm	TDK Electronics, ETD 29/16/10
$A_{L0}$ inductance factor of the ungapped set	2.2 µH	TDK Electronics, ETD 29/16/10
$W_A$ winding window area	97 mm²	TDK Electronics, ETD 29/16/10
$\mathrm{MLT}$ mean length per turn	52.8 mm	TDK Electronics, ETD 29/16/10
$N_\mathrm{min}$ fewest turns for Bmax, at Amin	38.59	Fewest turns that keep the peak flux density at a limit
$N_p$ primary turns	39	rounded up
$A_L$ inductance factor the design needs	394.5 nH	Inductance from the inductance factor
$\mu_e$ effective permeability of the ungapped set	1622	Inductance factor of a gapped core
$l_g$ gap, without fringing	0.1987 mm	Air-gap length that gives an inductance with N turns
$B_\mathrm{pk}$ peak flux density (at Amin)	296.9 mT	Peak flux density from inductance and peak current
$B_\mathrm{ac}$ ac flux-density amplitude	35.22 mT	AC flux-density amplitude from the current ripple
$N_s$ secondary turns	10	n Np, rounded
$\rho_w$ conductor resistivity (at Tw)	22.66 nΩ·m	Resistivity of a conductor at its temperature
$\delta$ skin depth (at fs)	0.2396 mm	Skin depth of a conductor
$A_w$ bare conductor area of one turn (primary)	0.1963 mm²	Copper area of one turn of round strands
$R_\mathrm{dc}$ winding dc resistance (primary)	237.7 mΩ	DC resistance of a winding
$\eta$ porosity of a layer (primary)	0.4664	Porosity of a layer of round conductors
$\varphi$ layer thickness over skin depth (primary)	1.263	Dowell's layer thickness for round conductors
$F_R$ ac resistance factor (at fs) (primary)	1.976	AC resistance factor of a foil winding (Dowell)
$P_\mathrm{dc}$ loss at the dc resistance (primary)	198.5 mW	Conduction loss in a resistance
$A_w$ bare conductor area of one turn (secondary)	0.6234 mm²	Copper area of one turn of round strands
$R_\mathrm{dc}$ winding dc resistance (secondary)	19.19 mΩ	DC resistance of a winding
$\eta$ porosity of a layer (secondary)	0.5877	Porosity of a layer of round conductors
$\varphi$ layer thickness over skin depth (secondary)	1.786	Dowell's layer thickness for round conductors
$F_R$ ac resistance factor (at fs) (secondary)	1.657	AC resistance factor of a foil winding (Dowell)
$P_\mathrm{dc}$ loss at the dc resistance (secondary)	184.4 mW	Conduction loss in a resistance
$K_u$ window utilization (both windings)	0.1432	Window utilization of a winding
$L_\mathrm{lk}$ leakage inductance (primary)	3.736 µH	Leakage inductance of a primary-secondary winding arrangement

Try P-S-P: the leakage falls to about a quarter. The flux swing is largest at the highest input voltage, where the duty ratio is lowest; enter that point’s ripple to read the core loss.

Limitations

• Round wire only. Foil and litz windings are not covered. Dowell’s factor assumes a one-dimensional field across the layers [Dowell 1966]. Near a gap the field is two-dimensional, and the factor does not describe the loss there.
• Fringing. The gap estimate ignores it (see above). For a table core, use the data sheet’s gapped sets.
• Flyback windings. The two windings conduct in turn, so their fields do not cancel as a transformer’s do. Each winding’s $F_R$ is taken with its own layers, from zero field on one side. The eddy-current loss that a winding’s field induces in the idle winding is not modelled.
• No core loss and no thermal model. The tool gives $B_\mathrm{ac}$ for the material’s loss data, and takes the winding temperature as entered.
• Core data. The ungapped set’s $A_L$ has a wide tolerance, +30/−20 % [TDK Electronics, E 25/13/7 (EF 25)], and the gap estimate follows it.

Screenshot

The magnetics designer with the buck inductor example.